The present invention generally relates to circuits that provide a temperature independent reference voltage, and more specifically, to bandgap voltage reference circuits.
In an integrated circuit, a bandgap reference circuit provides a substantially constant reference voltage output that is immune to variation in fabrication process, operating temperature, and supply voltage. The bandgap reference circuit makes use of the predictable behavior of bandgap energy of semiconductor material. A typical bandgap reference circuit employs a semiconductor bipolar pn junction (diode) device that has a negative temperature coefficient (i.e. its output voltage falls with rising temperature), and complements it with a pair of bipolar junction devices, each having a different emitter cross-sectional area, that generates a voltage difference that has a positive temperature coefficient, thereby producing a voltage output that is invariant to temperature change. FIG. 2 shows a typical voltage reference of the prior art. As shown, the reference circuit 10 could be viewed as having been constructed with three functional components: a proportional-to-absolute-temperature (PTAT) block 14 that provides a positive temperature coefficient, a diode connected bipolar junction transistor 16 that provides a negative temperature coefficient, and a current mirror 12 that joins PTAT block 14 and the bipolar junction transistor 16 together. The PTAT block 14 is made up of a first and second bipolar junction transistors 18 and 20 connected together at their bases, the first bipolar junction transistor 18 having an emitter cross-sectional area that is only a fraction of the second one. The current mirror block 12, which consists of NMOS transistors 26, 28, and 30, mirrors the current flowing though the PTAT block 14 to the diode connected bipolar junction transistor 32. Due to the differential in emitter cross-sectional area between the first bipolar junction transistor 18 and the second bipolar junction transistor 20, the current density going through each transistor differs, which gives rise to the effect that each transistor would have a different base-to-emitter voltage (Vbe). The difference between the respective base-to-emitter voltages, denoted as xcex94Vbe, can be derived by one skilled in the art to be:       Δ    ⁢          xe2x80x83        ⁢          V      be        =            kT      q        ⁢    ln    ⁢          xe2x80x83        ⁢    X  
where k is the Boltzmann""s constant, T is the absolute temperature, q is the electrical charge, and X is the scaling factor of the emitter cross-sectional area. As shown in the equation above, the term xcex94Vbe is directly proportional to the absolute temperature T. The reference current I 36 can then be expressed as   I  =                    V        R1            R1        =                            Δ          ⁢                      xe2x80x83                    ⁢                      V            be                          R1            =                        kT          ⁢                      xe2x80x83                    ⁢          ln          ⁢                      xe2x80x83                    ⁢          X                qR1            
Since the current I 36 is also mirrored to the branch with the diode connected bipolar junction transistor 32, the output reference voltage 34 can be expressed as       V    ref    =                    V        be            +              IR        2              =                  V        be            +                                    R            2                                R            1                          ⁢                              kT            ⁢                          xe2x80x83                        ⁢            ln            ⁢                          xe2x80x83                        ⁢            X                    q                    
As it is shown in the equation above, the reference voltage Vref 34 is a function of the Vbe of the diode connected bipolar junction transistor 32 and the xcex94Vbe of the first and second bipolar junction transistors 18 and 20, scaled by the ratio of R2 and R1.
A more robust prior art bandgap voltage reference circuit, which is shown in FIG. 3, employs an operational amplifier 40 to take the place of the current mirror 12. The op amp 40 provides a feedback control loop pathway that keeps the two input nodes of the amplifier 40 at approximately the same voltage in the steady state. In so doing, the voltage difference xcex94Vbe between the two diodes, D1 and D2, is amplified, which contributes to its higher accuracy. However, the higher accuracy comes with penalties in the form of added circuit complexity and increased power consumption as a typical op amp requires a biasing circuit that draws additional power and takes up additional space. It is the object of the present invention to have a bandgap voltage reference circuit that gets the benefit of having an op amp while at the same time does not substantially increase the circuit complexity and power consumption.
The above object of the present invention has been achieved by a bandgap voltage reference circuit that incorporates a unique 2-stage transconductance amplifier into a feedback control loop to improve the reference voltage accuracy and stability without the need for a biasing circuit. Having a high gain circuit gives the present invention a good power supply rejection ratio, and, contributes to its higher accuracy and stability. The elimination of the bias circuit provides the present invention with low power consumption and less circuit complexity.